LTC3780 [Linear Systems]
High Effi ciency, Synchronous, 4-Switch Buck-Boost Controller; 高艾菲效率，同步，四开关降压 - 升压型控制器
| 型号： | LTC3780 |
| 厂家： | 
Linear Systems |
| 描述： | High Effi ciency, Synchronous, 4-Switch Buck-Boost Controller |
| 文件： | 总28页 (文件大小：383K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC3780
High Efficiency, Synchronous,
4-Switch Buck-Boost Controller
FEATURES
DESCRIPTION
The LTC®3780 is a high performance buck-boost switch-
ing regulator controller that operates from input voltages
above, below or equal to the output voltage. The constant
frequency current mode architecture allows a phase-lock-
able frequency of up to 400kHz. With a wide 4V to 30V
(36V maximum) input and output range and seamless
transfers between operating modes, the LTC3780 is ideal
for automotive, telecom and battery-powered systems.
n
Single Inductor Architecture Allows V Above,
IN
Below or Equal to V
OUT
n
Wide V Range: 4V to 36V Operation
IN
n
Synchronous Rectification: Up to 98% Efficiency
n
Current Mode Control
n
±1% Output Voltage Accuracy: 0.8V < V
< 30V
OUT
n
n
n
n
n
n
n
n
n
n
Phase-Lockable Fixed Frequency: 200kHz to 400kHz
Power Good Output Voltage Monitor
Internal LDO for MOSFET Supply
Theoperatingmodeofthecontrollerisdeterminedthrough
the FCB pin. For boost operation, the FCB mode pin can
selectamongBurstMode® operation,discontinuousmode
and forced continuous mode. During buck operation, the
FCB mode pin can select among skip-cycle mode, discon-
tinuous mode and forced continuous mode. Burst Mode
operation and skip-cycle mode provide high efficiency
operation at light loads while forced continuous mode and
discontinuous mode operate at a constant frequency.
Quad N-Channel MOSFET Synchronous Drive
V
Disconnected from V During Shutdown
OUT
IN
Adjustable Soft-Start Current Ramping
Foldback Output Current Limiting
Selectable Low Current Modes
Output Overvoltage Protection
Available in 24-Lead SSOP and Exposed Pad
(5mm × 5mm) 32-Lead QFN Packages
Fault protection is provided by an output overvoltage
comparatorandinternalfoldbackcurrentlimiting.Apower
good output pin indicates when the output is within 7.5%
of its designed set point.
, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents including 5481178, 6304066, 5929620, 5408150, 6580258,
patent pending on current mode architecture and protection
APPLICATIONS
n
Automotive Systems
n
Telecom Systems
n
DC Power Distribution Systems
High Power Battery-Operated Devices
n
n
Industrial Control
TYPICAL APPLICATION
High Efficiency Buck-Boost Converter
V
12V
5A
OUT
V
IN
5V TO 32V
Efficiency and Power Loss
VOUT = 12V, ILOAD = 5A
100μF
16V
CER
+
22μF
50V
CER
+
330μF
16V
1μF
CER
4.7μF
V
IN
PGOOD INTV
A
B
D
C
CC
100
95
10
9
8
7
6
5
4
3
2
1
0
TG2
TG1
0.1μF
0.1μF
BOOST2
SW2
BOOST1
SW1
90
LTC3780
BG2
BG1
PLLIN
RUN
85
I
TH
105k
1%
2200pF
20k
ON/OFF
SS
V
0.1μF
OSENSE
FCB
80
75
70
SGND
SENSE SENSE PGND
7.5k
1%
+
–
0
5
10
15
V
20
25
30
35
0.010Ω
4.7μH
(V)
IN
3780 TA01b
3780 TA01
3780fe
1
LTC3780
ABSOLUTE MAXIMUM RATINGS (Note 1)
Input Supply Voltage (V )........................ –0.3V to 36V
Peak Output Current <10μs (TG1, TG2, BG1, BG2).....3A
INTV Peak Output Current ................................. 40mA
IN
Topside Driver Voltages
CC
(BOOST1, BOOST2) .................................. –0.3V to 42V
Switch Voltage (SW1, SW2) ........................ –5V to 36V
Operating Junction Temperature Range (Notes 5, 2, 7)
LTC3780E............................................. –40°C to 85°C
LTC3780I............................................ –40°C to 125°C
LTC3780MP ....................................... –55°C to 125°C
Junction Temperature (Note 2) ............................ 125°C
Storage Temperature Range................... –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
INTV , EXTV , (BOOST – SW1),
CC
CC
(BOOST2 – SW2), PGOOD.......................... –0.3V to 7V
RUN, SS....................................................... –0.3V to 6V
PLLIN Voltage.......................................... –0.3V to 5.5V
PLLFLTR Voltage....................................... –0.3V to 2.7V
FCB, STBYMD Voltages........................ –0.3V to INTV
TH OSENSE
SSOP Only........................................................ 300°C
CC
I , V
Voltages .............................. –0.3V to 2.4V
PIN CONFIGURATION
TOP VIEW
TOP VIEW
1
BOOST1
TG1
24
23
22
21
20
19
18
17
16
15
14
13
PGOOD
32 31 30 29 28 27 26 25
+
2
SS
+
SENSE
SENSE
I
1
2
3
4
5
6
7
8
24 SW1
3
SW1
SENSE
–
–
23
V
IN
4
V
IN
SENSE
EXTV
INTV
22
TH
CC
5
EXTV
CC
I
TH
V
21
OSENSE
SGND
CC
6
INTV
CC
V
OSENSE
33
20 BG1
7
BG1
SGND
RUN
FCB
PGND
19
8
PGND
BG2
RUN
18 BG2
17 SW2
9
FCB
PLLFTR
10
SW2
PLLFLTR
9
10 11 12 13 14 15 16
11
TG2
PLLIN
12
BOOST2
STBYMD
G PACKAGE
24-LEAD PLASTIC SSOP
UH PACKAGE
32-LEAD (5mm s 5mm) PLASTIC QFN
T
= 125°C, θ = 130°C/W
JA
JMAX
T
= 125°C, θ = 34°C/W
JMAX JA
EXPOSED PAD (PIN 33) IS GND, MUST BE SOLDERED TO PCB
3780fe
2
LTC3780
ORDER INFORMATION
LEAD FREE FINISH
LTC3780EG#PBF
LTC3780IG#PBF
LTC3780EUH#PBF
LTC3780IUH#PBF
LEAD BASED FINISH
LTC3780EG
TAPE AND REEL
LTC3780EG#TRPBF
LTC3780IG#TRPBF
LTC3780EUH#TRPBF
LTC3780IUH#TRPBF
TAPE AND REEL
LTC3780EG#TR
PART MARKING
LTC3780EG
LTC3780IG
3780
PACKAGE DESCRIPTION
TEMPERATURE RANGE
–40°C to 85°C
24-Lead Plastic SSOP
24-Lead Plastic SSOP
–40°C to 125°C
–40°C to 85°C
32-Lead (5mm × 5mm) Plastic QFN
32-Lead (5mm × 5mm) Plastic QFN
PACKAGE DESCRIPTION
3780I
–40°C to 125°C
TEMPERATURE RANGE
–40°C to 85°C
PART MARKING
LTC3780EG
LTC3780IG
LTC3780MPG
3780
24-Lead Plastic SSOP
LTC3780IG
LTC3780IG#TR
24-Lead Plastic SSOP
–40°C to 125°C
–55°C to 125°C
–40°C to 85°C
LTC3780MPG
LTC3780MPG#TR
LTC3780EUH#TR
LTC3780IUH#TR
24-Lead Plastic SSOP
LTC3780EUH
32-Lead (5mm × 5mm) Plastic QFN
32-Lead (5mm × 5mm) Plastic QFN
LTC3780IUH
3780I
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
l
l
V
Feedback Reference Voltage
I
= 1.2V, –40°C ≤ T ≤ 85°C (Note 3)
0.792
0.792
0.800
0.800
0.808
0.811
V
V
OSENSE
TH
–55°C ≤ T ≤ 125°C
I
Feedback Pin Input Current
(Note 3)
–5
–50
nA
VOSENSE
V
Output Voltage Load Regulation
(Note 3)
LOADREG
l
l
∆I = 1.2V to 0.7V
0.1
–0.1
0.5
–0.5
%
%
TH
∆I = 1.2V to 1.8V
TH
V
Reference Voltage Line Regulation
Error Amplifier Transconductance
Error Amplifier GBW
V
= 4V to 30V, I = 1.2V (Note 3)
0.002
0.32
0.6
0.02
%/V
mS
REF(LINEREG)
m(EA)
IN
TH
g
g
I
TH
= 1.2V, Sink/Source = 3μA (Note 3)
(Note 8)
(Note 4)
MHz
m(GBW)
I
Input DC Supply Current
Normal
Q
2400
1500
55
μA
μA
μA
Standby
V
V
= 0V, V
= 0V, V
> 2V
= Open
RUN
RUN
STBYMD
STBYMD
Shutdown Supply Current
70
V
Forced Continuous Threshold
Forced Continuous Pin Current
0.76
0.800
–0.18
5.3
0.84
–0.1
5.5
V
μA
V
FCB
I
V
= 0.85V
–0.30
FCB
FCB
V
Burst Inhibit (Constant Frequency)
Threshold
Measured at FCB Pin
BINHIBIT
l
UVLO
Undervoltage Reset
V
Falling
3.8
0.86
–380
0.7
4
V
V
IN
V
Feedback Overvoltage Lockout
Sense Pins Total Source Current
Start-Up Threshold
Measured at V
Pin
0.84
0.4
0.88
OVL
OSENSE
+
–
I
V
V
V
= V = 0V
SENSE
μA
V
SENSE
SENSE
V
V
Rising
STBYMD(START)
STBYMD(KA)
STBYMD
STBYMD
Keep-Alive Power-On Threshold
Rising, V
= 0V
1.25
V
RUN
3780fe
3
LTC3780
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
99
MAX
UNITS
%
DF MAX, Boost
DF MAX, Buck
Maximum Duty Factor
Maximum Duty Factor
RUN Pin On Threshold
Soft-Start Charge Current
Maximum Current Sense Threshold
% Switch C On
% Switch A On (in Dropout)
99
%
V
V
RUN
V
RUN
Rising
= 2V
1
1.5
1.2
2
V
RUN(ON)
I
0.5
μA
SS
l
l
V
Boost: V
= V – 50mV
REF
160
–110
185
–150
mV
mV
SENSE(MAX)
OSENSE
OSENSE
REF
Buck: V
= V – 50mV
–95
V
Minimum Current Sense Threshold
TG Rise Time
Discontinuous Mode
–6
50
45
45
55
80
mV
ns
ns
ns
ns
ns
SENSE(MIN,BUCK)
TG1, TG2 t
TG1, TG2 t
C
LOAD
C
LOAD
C
LOAD
C
LOAD
C
LOAD
= 3300pF (Note 5)
= 3300pF (Note 5)
= 3300pF (Note 5)
= 3300pF (Note 5)
= 3300pF Each Driver
r
TG Fall Time
f
BG1, BG2 t
BG1, BG2 t
BG Rise Time
r
BG Fall Time
f
TG1/BG1 t
BG1/TG1 t
TG2/BG2 t
BG2/TG2 t
Mode
TG1 Off to BG1 On Delay,
Switch C On Delay
1D
BG1 Off to TG1 On Delay,
Synchronous Switch D On Delay
C
LOAD
C
LOAD
C
LOAD
C
LOAD
C
LOAD
= 3300pF Each Driver
= 3300pF Each Driver
= 3300pF Each Driver
= 3300pF Each Driver
= 3300pF Each Driver
80
80
ns
ns
ns
ns
ns
ns
ns
2D
3D
4D
TG2 Off to BG2 On Delay,
Synchronous Switch B On Delay
BG2 Off to TG2 On Delay,
Switch A On Delay
80
BG1 Off to BG2 On Delay,
Switch A On Delay
250
250
200
180
Transition 1
Mode
Transition 2
BG2 Off to BG1 On Delay,
Synchronous Switch D On Delay
t
Minimum On-Time for Main Switch in
Boost Operation
Switch C (Note 6)
Switch B (Note 6)
ON(MIN,BOOST)
ON(MIN,BUCK)
t
Minimum On-Time for Synchronous
Switch in Buck Operation
Internal V Regulator
CC
l
l
V
Internal V Voltage
7V < V < 30V, V = 5V
EXTVCC
5.7
5.4
6
6.3
2
V
%
INTVCC
CC
IN
Internal V Load Regulation
I
CC
I
CC
= 0mA to 20mA, V = 5V
EXTVCC
0.2
5.7
300
150
∆V
CC
LDO(LOADREG)
V
EXTV Switchover Voltage
= 20mA, V
Rising
V
EXTVCC
CC
EXTVCC
EXTVCC
EXTV Switchover Hysteresis
mV
mV
∆V
∆V
CC
EXTVCC(HYS)
EXTV Switch Drop Voltage
I
CC
= 20mA, V
= 6V
300
CC
EXTVCC
Oscillator and Phase-Locked Loop
f
f
f
Nominal Frequency
V
V
V
= 1.2V
= 0V
260
170
340
300
200
400
50
330
220
440
kHz
kHz
kHz
kΩ
NOM
LOW
HIGH
PLLFLTR
PLLFLTR
PLLFLTR
Lowest Frequency
Highest Frequency
= 2.4V
R
PLLIN Input Resistance
Phase Detector Output Current
PLLIN
I
f
f
< f
> f
–15
15
μA
μA
PLLLPF
PLLIN
PLLIN
OSC
OSC
(Note 9)
3780fe
4
LTC3780
ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
PGOOD Output
PGOOD Upper Threshold
PGOOD Lower Threshold
PGOOD Hysteresis
V
V
V
Rising
5.5
7.5
–7.5
2.5
10
%
%
%
V
∆V
OSENSE
OSENSE
OSENSE
FBH
Falling
–5.5
–10
∆V
FBL
Returning
∆V
FB(HYST)
V
PGOOD Low Voltage
I
= 2mA
= 5V
0.1
0.3
1
PGL
PGOOD
I
PGOOD Leakage Current
V
μA
PGOOD
PGOOD
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 6: The minimum on-time condition is specified for an inductor
peak-to-peak ripple current ≥ 40% of I (see minimum on-time
considerations in the Applications Information section).
Note 7: The LTC3780E is guaranteed to meet performance specifications
from 0°C to 85°C. Performance over the –40°C to 85°C operating junction
temperature range is assured by design, characterization and correlation
with statistical process controls. The LTC3780I is guaranteed over the
–40°C to 125°C operating junction temperature range. The LTC3780MP
is guaranteed and tested over the full –55 to 125°C operating junction
temperature range.
MAX
Note 2: T for the QFN package is calculated from the temperature T and
J
A
power dissipation P according to the following formula:
D
T = T + (P • 34°C/W)
J
A
D
Note 3: The IC is tested in a feedback loop that servos V to a specified
ITH
voltage and measures the resultant V
.
OSENSE
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: Rise and fall times are measured using 10% and 90% levels. Delay
Note 8: This parameter is guaranteed by design.
Note 9: f
is the running frequency for the application.
OSC
times are measured using 50% levels.
3780fe
5
LTC3780
TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted.
Efficiency vs Output Current
(Boost Operation)
Efficiency vs Output Current
Efficiency vs Output Current
(Buck Operation)
100
90
80
70
60
50
40
100
90
80
70
60
50
40
100
90
80
70
60
50
40
BURST
BURST
DCM
SC
DCM
CCM
CCM
DCM
CCM
V
V
= 12V
V
IN
V
OUT
= 18V
V
V
= 6V
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
0.01
0.1
1
10
0.01
0.1
1
10
0.01
0.1
1
10
I
(A)
I
(A)
LOAD
I
(A)
LOAD
LOAD
3780 G02
3780 G03
3780 G01
Supply Current vs Input Voltage
Internal 6V LDO Line Regulation
EXTVCC Voltage Drop
2500
2000
1500
1000
500
6.5
6.0
5.5
5.0
120
100
V
= 0V
FCB
STANDBY
80
60
4.5
4.0
3.5
40
20
0
SHUTDOWN
0
0
5
10
15
20
25
30
35
20
INPUT VOLTAGE (V)
30
35
0
5
10
15
25
0
10
20
30
40
50
INPUT VOLTAGE (V)
CURRENT (mA)
3780 G04
3780 G05
3780 G06
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Switch Resistance
vs Temperature
Load Regulation
6.05
6.00
5.95
5.90
5.85
5.80
5.75
5.70
5.65
5.60
5.55
5
4
3
2
1
0
0
–0.1
–0.2
–0.3
–0.4
–0.5
V
IN
= 18V
INTV VOLTAGE
CC
V
IN
= 12V
V
IN
= 6V
EXTV SWITCHOVER THRESHOLD
CC
FCB = 0V
V
= 12V
OUT
–75 –50
0
25 50 75 100 125
–75 –50 –25
0
25 50 75 100 125
–25
0
1
2
3
4
5
TEMPERATURE (°C)
TEMPERATURE (°C)
LOAD CURRENT (A)
3780 G07
3780 G08
3780 G09
3780fe
6
LTC3780
TA = 25°C, unless otherwise noted.
TYPICAL PERFORMANCE CHARACTERISTICS
Continuous Current Mode
(CCM, VIN = 6V, VOUT = 12V)
Continuous Current Mode
(CCM, VIN = 12V, VOUT = 12V)
Continuous Current Mode
(CCM, VIN = 18V, VOUT = 12V)
SW2
10V/DIV
SW2
10V/DIV
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
V
OUT
V
OUT
V
100mV/DIV
OUT
100mV/DIV
100mV/DIV
I
I
I
L
L
L
2A/DIV
2A/DIV
2A/DIV
3780 G10
3780 G11
3780 G12
V
V
= 6V
5μs/DIV
V
V
= 12V
5μs/DIV
IN
OUT
V
V
= 18V
5μs/DIV
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
Burst Mode Operation
(VIN = 6V, VOUT = 12V)
Burst Mode Operation
(VIN = 12V, VOUT = 12V)
Skip-Cycle Mode
(VIN = 18V, VOUT = 12V)
SW2
10V/DIV
SW2
10V/DIV
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
V
V
OUT
OUT
V
OUT
500mV/DIV
200mV/DIV
100mV/DIV
I
L
I
L
2A/DIV
2A/DIV
I
L
1A/DIV
3780 G14
3780 G15
3780 G13
V
V
= 12V
10μs/DIV
V
V
= 18V
2.5μs/DIV
V
V
= 6V
25μs/DIV
IN
OUT
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
Discontinuous Current Mode
(DCM, VIN = 6V, VOUT = 12V)
Discontinuous Current Mode
(DCM, VIN = 12V, VOUT = 12V)
Discontinuous Current Mode
(DCM, VIN = 18V, VOUT = 12V)
SW2
10V/DIV
SW2
10V/DIV
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
V
V
OUT
OUT
V
OUT
100mV/DIV
100mV/DIV
100mV/DIV
I
L
1A/DIV
I
I
L
L
1A/DIV
2A/DIV
3780 G17
3780 G18
3780 G16
V
V
= 12V
5μs/DIV
V
V
= 18V
2.5μs/DIV
V
V
= 6V
5μs/DIV
IN
OUT
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
3780fe
7
LTC3780
TA = 25°C, unless otherwise noted.
TYPICAL PERFORMANCE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Undervoltage Reset
vs Temperature
Minimum Current Sense
Threshold vs Duty Factor (Buck)
–20
4.2
4.0
3.8
3.6
3.4
3.2
3.0
450
400
350
300
250
200
150
100
50
V
V
= 2.4V
= 1.2V
PLLFLTR
PLLFLTR
–40
–60
–80
V
= 0V
PLLFLTR
0
–75 –50 –25
0
25 50 75 100 125
100
80
60
40
20
0
–75 –50 –25
0
25
125
50 75 100
TEMPERATURE (°C)
DUTY FACTOR (%)
TEMPERATURE (°C)
3780 G20
3780 G21
3780 G19
Maximum Current Sense
Threshold vs Duty Factor (Boost)
Maximum Current Sense
Threshold vs Duty Factor (Buck)
Minimum Current Sense
Threshold vs Temperature
200
150
180
160
140
120
100
140
130
120
110
BOOST
100
50
0
–50
–100
BUCK
–150
50
TEMPERATURE (°C)
100 125
–75 –50 –25
0
25
75
0
20
40
60
80
100
0
20
40
60
80
100
DUTY FACTOR (%)
DUTY FACTOR (%)
3780 G24
3780 G22
3780 G23
Peak Current Threshold
vs VITH (Boost)
Valley Current Threshold
vs VITH (Buck)
Current Foldback Limit
200
150
100
50
100
50
200
160
120
80
BOOST
BUCK
0
–50
0
–100
–150
40
0
–50
–100
0
0.8
1.2
(V)
1.6
1.8
2.4
0
0.8
1.2
(V)
1.6
2.0
2.4
0.4
0.4
0
0.2
0.4
0.6
0.8
V
V
V
(V)
ITH
ITH
OSENSE
3780 G32
3780 G25
3780 G26
3780fe
8
LTC3780
TA = 25°C, unless otherwise noted.
TYPICAL PERFORMANCE CHARACTERISTICS
Load Step
Load Step
Load Step
V
V
V
OUT
500mV/DIV
OUT
OUT
500mV/DIV
500mV/DIV
I
L
I
5A/DIV
I
L
L
5A/DIV
5A/DIV
3780 G28
3780 G29
3780 G27
V
V
= 12V
200μs/DIV
V
V
= 6V
200μs/DIV
V
V
= 18V
200μs/DIV
IN
OUT
IN
OUT
IN
OUT
= 12V
= 12V
= 12V
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
Line Transient
Line Transient
V
IN
V
IN
10V/DIV
10V/DIV
V
OUT
V
OUT
500mV/DIV
500mV/DIV
I
I
L
L
1A/DIV
1A/DIV
3780 G30
3780 G31
V
I
IN
= 12V
= 1A
500μs/DIV
V
I
= 12V
= 1A
500μs/DIV
OUT
LOAD
OUT
LOAD
IN
V
STEP: 7V TO 20V
V
STEP: 20V TO 7V
CONTINUOUS MODE
CONTINUOUS MODE
PIN FUNCTIONS (SSOP/QFN)
PGOOD (Pin 1/Pin 30): Open-Drain Logic Output. PGOOD
is pulled to ground when the output voltage is not within
7.5% of the regulation point.
–
+
voltage and built-in offsets between SENSE and SENSE
pins, in conjunction with R
threshold.
, set the current trip
SENSE
–
SS (Pin 2/Pin 31): Soft-start reduces the input power
sources’ surge currents by gradually increasing the
controller’s current limit. A minimum value of 6.8nF is
recommended on this pin.
SENSE (Pin 4/Pin 2): The (–) Input to the Current Sense
and Reverse Current Detect Comparators.
I
(Pin 5/Pin 3): Current Control Threshold and Error
TH
Amplifier Compensation Point. The current comparator
threshold increases with this control voltage. The voltage
ranges from 0V to 2.4V.
+
SENSE (Pin 3/Pin 1): The (+) Input to the Current Sense
and Reverse Current Detect Comparators. The I pin
TH
3780fe
9
LTC3780
PIN FUNCTIONS (SSOP/QFN)
V
(Pin 6/Pin 4): Error Amplifier Feedback Input.
BOOST2, BOOST1 (Pins 13, 24/Pins 14, 27): Boosted
OSENSE
This pin connects the error amplifier input to an external
Floating Driver Supply. The (+) terminal of the bootstrap
resistor divider from V
.
capacitorC andC (Figure11)connectshere.TheBOOST2
OUT
A B
pin swings from a diode voltage below INTV up to V
CC
IN
SGND (Pin 7/Pin 5): Signal Ground. All small-signal com-
ponents and compensation components should connect
to this ground, which should be connected to PGND at a
single point.
+ INTV . The BOOST1 pin swings from a diode voltage
CC
below INTV up to V
+ INTV .
CC
OUT
CC
TG2,TG1(Pins14,23/Pins15,26):TopGateDrive.Drives
the top N-channel MOSFET with a voltage swing equal to
RUN (Pin 8/Pin 6): Run Control Input. Forcing the RUN
pin below 1.5V causes the IC to shut down the switching
regulatorcircuitry.Thereisa100kresistorbetweentheRUN
pin and SGND in the IC. Do not apply >6V to this pin.
INTV superimposed on the switch node voltage SW.
CC
SW2,SW1(Pins15,22/Pins17,24):SwitchNode.The(–)
terminal of the bootstrap capacitor C and C (Figure 11)
A
B
connectshere. TheSW2pinswingsfromaSchottkydiode
FCB (Pin 9/Pin 7): Forced Continuous Control Input. The
voltage applied to this pin sets the operating mode of the
controller. When the applied voltage is less than 0.8V, the
forced continuous current mode is active. When this pin
is allowed to float, the Burst Mode operation is active in
boost operation and the skip-cycle mode is active in buck
(external) voltage drop below ground up to V . The SW1
IN
pin swings from a Schottky diode (external) voltage drop
below ground up to V
.
OUT
BG2, BG1 (Pins 16, 18/Pins 18, 20): Bottom Gate Drive.
Drives the gate of the bottom N-channel MOSFET between
operation. When the pin is tied to INTV , the constant
ground and INTV .
CC
CC
frequency discontinuous current mode is active in buck
PGND (Pin 17/Pin 19): Power Ground. Connect this pin
or boost operation.
closelytothesourceofthebottomN-channelMOSFET,the
PLLFLTR (Pin 10/Pin 8): The phase-locked loop’s
lowpass filter is tied to this pin. Alternatively, this pin can
be driven with an AC or DC voltage source to vary the
frequency of the internal oscillator.
(–)terminalofC andthe(–)terminalofC (Figure11).
VCC IN
INTV (Pin19/Pin21):Internal6VRegulatorOutput. The
CC
driver and control circuits are powered from this voltage.
Bypass this pin to ground with a minimum of 4.7μF low
ESR tantalum or ceramic capacitor.
PLLIN (Pin 11/Pin 10): External Synchronization Input to
Phase Detector. This pin is internally terminated to SGND
with 50kΩ. The phase-locked loop will force the rising
bottom gate signal of the controller to be synchronized
with the rising edge of the PLLIN signal.
EXTV (Pin20/Pin22):ExternalV Input.WhenEXTV
CC
CC
CC
CC
exceeds5.7V,aninternalswitchconnectsthispintoINTV
andshutsdowntheinternalregulatorsothatthecontroller
andgatedrivepowerisdrawnfromEXTV .Donotexceed
CC
STBYMD (Pin 12/Pin 11): LDO Control Pin. Determines
whethertheinternalLDOremainsactivewhenthecontrol-
ler is shut down. See Operation section for details. If the
STBYMD pin is pulled to ground, the SS pin is internally
pulled to ground, preventing start-up and thereby provid-
ing a single control pin for turning off the controller. To
keep the LDO active when RUN is low, for example to
power a “wake up” circuit which controls the state of the
RUN pin, bypass STBYMD to signal ground with a 0.1μF
7V at this pin and ensure that EXTV < V .
CC IN
V (Pin 21/Pin 23): Main Input Supply. Bypass this pin
IN
to SGND with an RC filter (1Ω, 0.1μF).
Exposed Pad (Pin 33, QFN Only): This pin is SGND and
must be soldered to PCB ground.
capacitor, or use a resistor divider from V to keep the
pin within 2V to 5V.
IN
3780fe
10
LTC3780
BLOCK DIAGRAM
INTV
V
IN
CC
BOOST2
TG2
STBYMD
FCB
FCB
I
LIM
SW2
+
–
BUCK
LOGIC
INTV
CC
BG2
R
SENSE
PGND
BG1
I
REV
+
–
FCB
INTV
CC
BOOST
LOGIC
SW1
TG1
1.2V
4(V
)
FB
I
CMP
+
–
BOOST1
0.86V
1.2μA
OV
EA
SS
–
+
INTV
CC
V
OUT
RUN
SLOPE
V
OSENSE
100k
–
+
V
FB
0.80V
I
TH
SHDN
RST
FB
RUN/
SS
4(V
)
+
SENSE
–
SENSE
PLLIN
50k
V
REF
F
PHASE DET
IN
V
IN
V
IN
+
–
5.7V
PLLFLTR
CLK
R
LP
OSCILLATOR
6V
LDO
REG
C
LP
EXTV
INTV
CC
–
+
0.86V
6V
+
CC
PGOOD
INTERNAL
SUPPLY
SGND
V
OSENSE
–
+
0.74V
3780 BD
3780fe
11
LTC3780
OPERATION
MAIN CONTROL LOOP
V
V
OUT
IN
The LTC3780 is a current mode controller that provides an
output voltage above, equal to or below the input voltage.
TheLTCproprietarytopologyandcontrolarchitectureem-
ploys a current-sensing resistor in buck or boost modes.
The sensed inductor current is controlled by the voltage
TG2
BG2
A
D
TG1
BG1
L
SW2
SW1
B
C
R
SENSE
on the I pin, which is the output of the amplifier EA. The
TH
3780 F01
V
pin receives the voltage feedback signal, which is
OSENSE
Figure 1. Simplified Diagram of the Output Switches
compared to the internal reference voltage by the EA.
The top MOSFET drivers are biased from floating boost-
strapcapacitorsC andC (Figure11), whicharenormally
98%
MAX
BOOST
D
A
B
A ON, B OFF
rechargedthroughanexternaldiodewhenthetopMOSFET
is turned off. Schottky diodes across the synchronous
switch D and synchronous switch B are not required, but
provide a lower drop during the dead time. The addition of
the Schottky diodes will typically improve peak efficiency
by 1% to 2% at 400kHz.
BOOST REGION
PWM C, D SWITCHES
D
MIN
BOOST
FOUR SWITCH PWM
BUCK/BOOST REGION
BUCK REGION
D
MAX
BUCK
D ON, C OFF
PWM A, B SWITCHES
3%
MIN
BUCK
D
3780 F02
The main control loop is shut down by pulling the RUN
pin low. When the RUN pin voltage is higher than 1.5V, an
internal 1.2μA current source charges soft-start capacitor
Figure 2. Operating Mode vs Duty Cycle
C
at the SS pin. The I voltage is then clamped to the
SS
TH
and switch A is turned on for the remainder of the cycle.
switches A and B will alternate, behaving like a typical
synchronous buck regulator. The duty cycle of switch A
increases until the maximum duty cycle of the converter
SS voltage while C is slowly charged during start-up.
SS
This “soft-start” clamping prevents abrupt current from
being drawn from the input power supply.
in buck mode reaches D , given by:
MAX_BUCK
POWER SWITCH CONTROL
D
= 100% – D
BUCK-BOOST
MAX_BUCK
Figure 1 shows a simplified diagram of how the four
where D
range:
= duty cycle of the buck-boost switch
BUCK-BOOST
power switches are connected to the inductor, V , V
IN OUT
and GND. Figure 2 shows the regions of operation for the
LTC3780asafunctionofdutycycleD. Thepowerswitches
are properly controlled so the transfer between modes is
D
= (200ns • f) • 100%
BUCK-BOOST
and f is the operating frequency in Hz.
continuous. When V approaches V , the buck-boost
IN
OUT
Figure 3 shows typical buck mode waveforms. If V
region is reached; the mode-to-mode transition time is
IN
approaches V , the buck-boost region is reached.
typically 200ns.
OUT
Buck-Boost (V ≅ V
)
Buck Region (V > V
)
IN
OUT
IN
OUT
When V is close to V , the controller is in buck-boost
Switch D is always on and switch C is always off during
this mode. At the start of every cycle, synchronous switch
B is turned on first. Inductor current is sensed when
synchronous switch B is turned on. After the sensed in-
ductor current falls below the reference voltage, which is
IN
OUT
mode. Figure 4 shows typical waveforms in this mode.
Every cycle, if the controller starts with switches B and D
turned on, switches A and C are then turned on. Finally,
switches A and D are turned on for the remainder of the
time. If the controller starts with switches A and C turned
proportional to V , synchronous switch B is turned off
ITH
3780fe
12
LTC3780
OPERATION
the remainder of the cycle. switches C and D will alternate,
behaving like a typical synchronous boost regulator.
CLOCK
SWITCH A
SWITCH B
ThedutycycleofswitchCdecreasesuntiltheminimumduty
cycle of the converter in boost mode reaches D
given by:
,
MIN_BOOST
0V
SWITCH C
SWITCH D
HIGH
D
= D
MIN_BOOST
BUCK-BOOST
I
L
where D
is the duty cycle of the buck-boost
3780 F03
BUCK-BOOST
switch range:
Figure 3. Buck Mode (VIN > VOUT
)
D
= (200ns • f) • 100%
BUCK-BOOST
and f is the operating frequency in Hz.
CLOCK
Figure 5 shows typical boost mode waveforms. If V ap-
SWITCH A
IN
proaches V , the buck-boost region is reached.
OUT
SWITCH B
SWITCH C
SWITCH D
CLOCK
HIGH
0V
SWITCH A
SWITCH B
I
L
3780 F04a
SWITCH C
SWITCH D
(4a) Buck-Boost Mode (VIN ≥ VOUT
)
I
L
CLOCK
3780 F05
SWITCH A
SWITCH B
Figure 5. Boost Mode (VIN < VOUT
)
LOW CURRENT OPERATION
SWITCH C
SWITCH D
The FCB pin is used to select among three modes for both
buck and boost operations by accepting a logic input.
Figure 6 shows the different modes.
I
L
3780 F04b
FCB PIN
0V to 0.75V
0.85V to 5V
>5.3V
BUCK MODE
BOOST MODE
(4b) Buck-Boost Mode (VIN ≤ VOUT
)
Force Continuous Mode
Skip-Cycle Mode
Force Continuous Mode
Burst Mode Operation
DCM with Constant Freq
Figure 4. Buck-Boost Mode
DCM with Constant Freq
on, switches B and D are then turned on. Finally, switches
A and D are turned on for the remainder of the time.
Figure 6. Different Operating Modes
When the FCB pin voltage is lower than 0.8V, the controller
behavesasacontinuous,PWMcurrentmodesynchronous
switching regulator. In boost mode, switch A is always on.
switch C and synchronous switch D are alternately turned
on to maintain the output voltage independent of direction
of inductor current. Every ten cycles, switch A is forced off
Boost Region (V < V
)
IN
OUT
Switch A is always on and synchronous switch B is always
off in boost mode. Every cycle, switch C is turned on first.
Inductor current is sensed when synchronous switch C is
turned on. After the sensed inductor current exceeds the
reference voltage which is proportional to V , switch C
is turned off and synchronous switch D is turned on for
for about 300ns to allow boost capacitor C (Figure 13) to
A
ITH
recharge. In buck mode, synchronous switch D is always
3780fe
13
LTC3780
OPERATION
on. switch A and synchronous switch B are alternately
turned on to maintain the output voltage independent of
direction of inductor current. Every ten cycles, synchro-
controller will enter continuous current buck mode for
one cycle to discharge inductor current. In the following
cycle, thecontrollerwillresumeDCMboostoperation. For
buckoperation,constantfrequencydiscontinuouscurrent
mode sets a minimum negative inductor current level.
synchronous switch B is turned off whenever inductor
current is lower than this level. At very light loads, this
constant frequency operation is not as efficient as Burst
Mode operation or skip-cycle, but does provide lower
noise, constant frequency operation.
nous switch D is forced off for about 300ns to allow C
B
to recharge. This is the least efficient operating mode at
light load, but may be desirable in certain applications. In
this mode, the output can source or sink current.
WhentheFCBpinvoltageisbelowV
–1V,butgreater
INTVCC
than 0.8V, the controller enters Burst Mode operation in
boost operation or enters skip-cycle mode in buck opera-
tion. During boost operation, Burst Mode operation sets a
minimum output current level before inhibiting the switch
C and turns off synchronous switch D when the inductor
current goes negative. This combination of requirements
FREQUENCY SYNCHRONIZATION AND
FREQUENCY SETUP
The phase-locked loop allows the internal oscillator to be
synchronized to an external source via the PLLIN pin. The
phase detector output at the PLLFLTR pin is also the DC
frequency control input of the oscillator. The frequency
ranges from 200kHz to 400kHz, corresponding to a DC
voltage input from 0V to 2.4V at PLLFLTR. When locked,
the PLL aligns the turn on of the top MOSFET to the ris-
ing edge of the synchronizing signal. When PLLIN is left
open, the PLLFLTR pin goes low, forcing the oscillator to
its minimum frequency.
will, at low currents, force the I pin below a voltage
TH
threshold that will temporarily inhibit turn-on of power
switches C and D until the output voltage drops. There is
100mV of hysteresis in the burst comparator tied to the
I
TH
pin. This hysteresis produces output signals to the
MOSFETs C and D that turn them on for several cycles,
followed by a variable “sleep” interval depending upon the
loadcurrent.Themaximumoutputvoltagerippleislimited
to 3% of the nominal DC output voltage as determined
by a resistive feedback divider. During buck operation at
no load, switch A is turned on for its minimum on-time.
This will not occur every clock cycle when the output load
current drops below 1% of the maximum designed load.
The body diode of synchronous switch B or the Schottky
diode, which is in parallel with switch B, is used to dis-
charge the inductor current; switch B only turns on every
INTV /EXTV Power
CC
CC
Power for all power MOSFET drivers and most inter-
nal circuitry is derived from the INTV pin. When the
CC
EXTV pin is left open, an internal 6V low dropout linear
CC
regulator supplies INTV power. If EXTV is taken above
CC
CC
5.7V, the 6V regulator is turned off and an internal switch
ten clock cycles to allow C to recharge. As load current
B
is turned on, connecting EXTV to INTV . This allows
CC
CC
is applied, switch A turns on every cycle, and its on-time
begins to increase. At higher current, switch B turns on
briefly after each turn-off of switch A. switches C and D
remain off at light load, except to refresh CA (Figure 11)
every 10 clock cycles. In Burst Mode operation/skip-cycle
mode, the output is prevented from sinking current.
the INTV power to be derived from a high efficiency
CC
external source.
POWER GOOD (PGOOD) PIN
ThePGOODpinisconnectedtoanopendrainofaninternal
MOSFET. TheMOSFETturnsonandpullsthepinlowwhen
the output is not within 7.5% of the nominal output level
as determined by the resistive feedback divider. When
the output meets the 7.5% requirement, the MOSFET
is turned off and the pin is allowed to be pulled up by an
external resistor to a source of up to 7V.
When the FCB pin voltage is tied to the INTV pin, the
CC
controllerentersconstantfrequencydiscontinuouscurrent
mode (DCM). For boost operation, synchronous switch D
is held off whenever the I pin is below a threshold volt-
TH
age. In every cycle, switch C is used to charge inductor
current. After the output voltage is high enough, the
3780fe
14
LTC3780
OPERATION
FOLDBACK CURRENT
SHORT-CIRCUIT PROTECTION AND CURRENT LIMIT
Foldback current limiting is activated when the output
voltage falls below 70% of its nominal level, reducing
power waste. During start-up, foldback current limiting
is disabled.
SwitchAon-timeislimitedbyoutputvoltage.Whenoutput
voltage is reduced and is lower than its nominal level,
switch A on-time will be reduced.
In every boost mode cycle, current is limited by a voltage
reference, which is proportional to the I pin voltage. The
TH
INPUT UNDERVOLTAGE RESET
maximum sensed current is limited to 160mV. In every
buck mode cycle, the maximum sensed current is limited
to 130mV.
The SS capacitor will be reset if the input voltage is al-
lowed to fall below approximately 4V. The SS capacitor
will attempt to charge through a normal soft-start ramp
after the input voltage rises above 4V.
STANDBY MODE PIN
TheSTBYMDpinisathree-stateinputthatcontrolscircuitry
within the IC as follows: When the STBYMD pin is held at
ground, the SS pin is pulled to ground. When the pin is
left open, the internal SS current source charges the SS
capacitor, allowing turn-on of the controller and activat-
ing necessary internal biasing. When the STBYMD pin is
taken above 2V, the internal linear regulator is turned on
independentofthestateontheRUNandSSpins,providing
an output power source for “wake-up” circuitry. Bypass
the pin with a small capacitor (0.1μF) to ground if the pin
is not connected to a DC potential.
OUTPUT OVERVOLTAGE PROTECTION
An overvoltage comparator guards against transient over-
shoots (>7.5%) as well as other more serious conditions
thatmayovervoltagetheoutput.Inthiscase,synchronous
switch B and synchronous switch D are turned on until the
overvoltage condition is cleared or the maximum negative
current limit is reached. When inductor current is lower
than the maximum negative current limit, synchronous
switch B and synchronous switch D are turned off, and
switch A and switch C are turned on until the inductor
current reaches another negative current limit. If the
comparator still detects an overvoltage condition, switch
A and switch C are turned off, and synchronous switch B
and synchronous switch D are turned on again.
3780fe
15
LTC3780
APPLICATIONS INFORMATION
Figure 11 is a basic LTC3780 application circuit. External
component selection is driven by the load requirement,
Inductor Selection
The operating frequency and inductor selection are inter-
relatedinthathigheroperatingfrequenciesallowtheuseof
smaller inductor and capacitor values. The inductor value
has a direct effect on ripple current. The inductor current
and begins with the selection of R
and the inductor
SENSE
value. Next, the power MOSFETs are selected. Finally, C
IN
and C
are selected. This circuit can be configured for
OUT
operation up to an input voltage of 36V.
ripple ∆I is typically set to 20% to 40% of the maximum
L
inductor current at boost mode V
. For a given ripple
IN(MIN)
Selection of Operation Frequency
the inductance terms in continuous mode are as follows:
The LTC3780 uses a constant frequency architecture and
has an internal voltage controlled oscillator. The switching
frequencyisdeterminedbytheinternaloscillatorcapacitor.
This internal capacitor is charged by a fixed current plus
an additional current that is proportional to the voltage
appliedtothePLLFLTRpin.Thefrequencyofthisoscillator
can be varied over a 2-to-1 range. The PLLFLTR pin can
be grounded to lower the frequency to 200kHz or tied to
2.4V to yield approximately 400kHz. When PLLIN is left
open, the PLLFLTR pin goes low, forcing the oscillator to
minimum frequency.
V
2 t V
o V
t100
(
)
IN(MIN)
OUT
IN(MIN)
LBOOST
>
H,
2
ƒ tIOUT(MAX) t ꢀRipple t VOUT
VOUT t VIN(MAX) o VOUT t100
(
)
LBUCK
>
H
ƒ tIOUT(MAX) t ꢀRipple t V
IN(MAX)
where:
f is operating frequency, Hz
% Ripple is allowable inductor current ripple, %
A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure 7. As the operating frequency
isincreasedthegatechargelosseswillbehigher, reducing
efficiency. The maximum switching frequency is approxi-
mately 400kHz.
V
V
V
I
is minimum input voltage, V
is maximum input voltage, V
is output voltage, V
IN(MIN)
IN(MAX)
OUT
is maximum output load current
OUT(MAX)
For high efficiency, choose an inductor with low core loss,
such as ferrite and molypermalloy (from Magnetics, Inc.).
Also,theinductorshouldhavelowDCresistancetoreduce
450
400
350
300
250
200
150
100
50
2
theI Rlosses,andmustbeabletohandlethepeakinductor
current without saturating. To minimize radiated noise,
use a toroid, pot core or shielded bobbin inductor.
R
SENSE
Selection and Maximum Output Current
R
SENSE
is chosen based on the required output current.
The current comparator threshold sets the peak of the
inductorcurrentinboostmodeandthemaximuminductor
valleycurrentinbuckmode. Inboostmode, themaximum
0
0
2
2.5
0.5
1
1.5
PLLFLTR PIN VOLTAGE (V)
3780 F07
average load current at V
is:
IN(MIN)
Figure 7. Frequency vs PLLFLTR Pin Voltage
V
⎛
⎞
ΔIL
2
160mV
IN(MIN)
IOUT(MAX,BOOST)
=
n
s
⎜
⎟
R
VOUT
⎝
⎠
SENSE
3780fe
16
LTC3780
APPLICATIONS INFORMATION
where ∆I is peak-to-peak inductor ripple current. In buck
to handle the maximum RMS current. For buck operation,
the input RMS current is given by:
L
mode, the maximum average load current is:
130mV ΔIL
VOUT
V
IN
V
IN
VOUT
IOUT(MAX,BUCK)
=
+
IRMS ≈IOUT(MAX)
•
•
– 1
RSENSE
2
Figure 8 shows how the load current (I
varies with input and output voltage
• R
)
MAXLOAD
SENSE
This formula has a maximum at V = 2V , where
IN
OUT
I
= I
/2. This simple worst-case condition
RMS
OUT(MAX)
is commonly used for design because even significant
deviations do not offer much relief. Note that ripple cur-
rentratingsfromcapacitormanufacturersareoftenbased
on only 2000 hours of life which makes it advisable to
derate the capacitor.
The maximum current sensing R
mode is:
value for the boost
SENSE
RSENSE(MAX)
=
2s160mV sV
IN(MIN)
2sIOUT(MAX,BOOST) sVOUT + ΔIL,BOOST sV
In boost mode, the discontinuous current shifts from the
IN(MIN)
input to the output, so C
must be capable of reducing
OUT
The maximum current sensing R
mode is:
value for the buck
the output voltage ripple. The effects of ESR (equivalent
series resistance) and the bulk capacitance must be
considered when choosing the right capacitor for a given
output ripple voltage. The steady ripple due to charging
and discharging the bulk capacitance is given by:
SENSE
2s130mV
2sIOUT(MAX,BUCK) – ΔIL,BUCK
RSENSE(MAX)
=
The final R
SENSE(MAX)
30% margin is usually recommended.
value should be lower than the calculated
SENSE
IOUT(MAX) • V
– V
IN(MIN)
OUT
(
)
Ripple(Boost,Cap) =
Ripple(Buck,Cap) =
V
V
R
in both the boost and buck modes. A 20% to
COUT • VOUT • f
IOUT(MAX) • VIN(MAX) – V
(
)
OUT
C and C Selection
IN
OUT
COUT • VIN(MAX) • f
In boost mode, input current is continuous. In buck mode,
inputcurrentisdiscontinuous.Inbuckmode,theselection
of input capacitor C is driven by the need to filter the
input square wave current. Use a low ESR capacitor sized
where C
is the output filter capacitor.
OUT
IN
The steady ripple due to the voltage drop across the ESR
is given by:
160
∆V
∆V
= I
• ESR
• ESR
BOOST,ESR
L(MAX,BOOST)
150
140
130
120
110
100
= I
BUCK,ESR
L(MAX,BUCK)
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount
packages. Ceramic capacitors have excellent low ESR
characteristics but can have a high voltage coefficient.
Capacitors are now available with low ESR and high ripple
current ratings, such as OS-CON and POSCAP.
0.1
1
10
V
/V
(V)
IN OUT
3780 F08
Figure 8. Load Current vs VIN/VOUT
3780fe
17
LTC3780
APPLICATIONS INFORMATION
Power MOSFET Selection and
Efficiency Considerations
Switch B operates in buck mode as the synchronous
rectifier. Its power dissipation at maximum output current
is given by:
The LTC3780 requires four external N-channel power
MOSFETs,twoforthetopswitches(switchAandD,shown
inFigure1)andtwoforthebottomswitches(switchBand C
shown in Figure 1). Important parameters for the power
V – VOUT
IN
P
=
sIOUT(MAX)2 s ρT sRDS(ON)
B,BUCK
V
IN
Switch C operates in boost mode as the control switch. Its
power dissipation at maximum current is given by:
MOSFETs are the breakdown voltage V
, threshold
BR,DSS
, reverse transfer
voltage V
, on-resistance R
GS,TH
DS(ON)
and maximum current I
capacitance C
.
RSS
DS(MAX)
The drive voltage is set by the 6V INTV supply. Con-
V
– V V
IN OUT
CC
(
)
OUT
PC,BOOST
=
sIOUT(MAX)2 s ρT sRDS(ON)
sequently, logic-level threshold MOSFETs must be used
in LTC3780 applications. If the input voltage is expected
to drop below 5V, then the sub-logic threshold MOSFETs
should be considered.
2
V
IN
IOUT(MAX)
+ k s VOUT3 s
sCRSS s f
V
IN
In order to select the power MOSFETs, the power dis-
sipated by the device must be known. For switch A, the
maximumpowerdissipationhappensinboostmode,when
it remains on all the time. Its maximum power dissipation
at maximum output current is given by:
whereC isusuallyspecifiedbytheMOSFETmanufactur-
RSS
ers. The constant k, which accounts for the loss caused
by reverse recovery current, is inversely proportional to
the gate drive current and has an empirical value of 1.7.
For switch D, the maximum power dissipation happens in
boost mode, when its duty cycle is higher than 50%. Its
maximum power dissipation at maximum output current
is given by:
2
⎛ VOUT
⎞
⎠
PA,BOOST
=
sIOUT(MAX) s ρT sRDS(ON)
⎜
⎟
⎝ V
IN
where ρ is a normalization factor (unity at 25°C) ac-
T
2
⎛
⎞
⎟
⎠
VOUT
V
VOUT
counting for the significant variation in on-resistance with
temperature,typicallyabout0.4%/°CasshowninFigure 9.
For a maximum junction temperature of 125°C, using a
IN
PD,BOOST
=
s
sIOUT(MAX) sρT sRDS(ON)
⎜
V
⎝
IN
For the same output voltage and current, switch A has the
highest power dissipation and switch B has the lowest
power dissipation unless a short occurs at the output.
value ρ = 1.5 is reasonable.
T
2.0
1.5
1.0
0.5
0
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
T = T + P • R
J
A
TH(JA)
The R
to be used in the equation normally includes
TH(JA)
the R
for the device plus the thermal resistance from
TH(JC)
the case to the ambient temperature (R
). This value
TH(JC)
of T can then be compared to the original, assumed value
J
50
100
–50
150
0
used in the iterative calculation process.
JUNCTION TEMPERATURE (°C)
3780 F09
Figure 9. Normalized RDS(ON) vs Temperature
3780fe
18
LTC3780
APPLICATIONS INFORMATION
Schottky Diode (D1, D2) Selection
and Light Load Operation
INTV Regulator
CC
An internal P-channel low dropout regulator produces 6V
at the INTV pin from the V supply pin. INTV powers
TheSchottkydiodesD1andD2showninFigure1conduct
during the dead time between the conduction of the power
MOSFET switches. They are intended to prevent the body
diode of synchronous switches B and D from turning on
and storing charge during the dead time. In particular, D2
significantly reduces reverse recovery current between
switch D turn-off and switch C turn-on, which improves
converter efficiency and reduces switch C voltage stress.
In order for the diode to be effective, the inductance
between it and the synchronous switch must be as small
as possible, mandating that these components be placed
adjacently.
CC
IN
CC
the drivers and internal circuitry within the LTC3780. The
INTV pin regulator can supply a peak current of 40mA
CC
and must be bypassed to ground with a minimum of 4.7μF
tantalum,10μFspecialpolymerorlowESRtypeelectrolytic
capacitor. A1μFceramiccapacitorplaceddirectlyadjacent
to the INTV and PGND IC pins is highly recommended.
CC
Good bypassing is necessary to supply the high transient
current required by MOSFET gate drivers.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi-
mum junction temperature rating for the LTC3780 to be
exceeded.Thesystemsupplycurrentisnormallydominated
by the gate charge current. Additional external loading of
In buck mode, when the FCB pin voltage is 0.85 < V
FCB
< 5V, the converter operates in skip-cycle mode. In this
mode, synchronous switch B remains off until the induc-
tor peak current exceeds one-fifth of its maximum peak
current. As a result, D1 should be rated for about one-half
to one-third of the full load current.
the INTV also needs to be taken into account for the
CC
power dissipation calculations. The total INTV current
CC
can be supplied by either the 6V internal linear regulator
or by the EXTV input pin. When the voltage applied to
CC
the EXTV pin is less than 5.7V, all of the INTV current
CC
CC
In boost mode, when the FCB pin voltage is higher than
5.3V,theconverteroperatesindiscontinuouscurrentmode.
In this mode, synchronous switch D remains off until the
inductor peak current exceeds one-fifth of its maximum
peak current. As a result, D2 should be rated for about
one-third to one-fourth of the full load current.
is supplied by the internal 6V linear regulator. Power dis-
sipation for the IC in this case is V • I , and overall
IN INTVCC
efficiency is lowered. The junction temperature can be
estimated by using the equations given in Note 2 of the
ElectricalCharacteristics.Forexample,atypicalapplication
operating in continuous current mode might draw 24mA
Inbuckmode,whentheFCBpinvoltageishigherthan5.3V,
the converter operates in constant frequency discontinu-
ous current mode. In this mode, synchronous switch B
remains on until the inductor valley current is lower than
the sense voltage representing the minimum negative
from a 24V supply when not using the EXTV pin:
CC
T = 70°C + 24mA • 24V • 34°C/W = 90°C
J
Use of the EXTV input pin reduces the junction tem-
CC
perature to:
inductor current level (V
= –5mV). Both switch A
SENSE
T = 70°C + 24mA • 6V • 34°C/W = 75°C
J
and B are off until next clock signal.
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum V .
In boost mode, when the FCB pin voltage is 0.85 < V
FCB
< 5.3V, the converter operates in Burst Mode operation.
In this mode, the controller clamps the peak inductor
current to approximately 20% of the maximum inductor
current. The output voltage ripple can increase during
Burst Mode operation.
IN
3780fe
19
LTC3780
APPLICATIONS INFORMATION
EXTV Connection
supply the gate drive voltage for the topside MOSFET
switches A and D. When the top MOSFET switch A turns
CC
The LTC3780 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTVCC rises above 5.7V, the
internal regulator is turned off and a switch connects the
EXTVCC pin to the INTVCC pin thereby supplying internal
power. The switch remains closed as long as the voltage
applied to EXTVCC remains above 5.5V. This allows the
MOSFET driver and control power to be derived from the
output when (5.7V < VOUT < 7V) and from the internal
regulator when the output is out of regulation (start-up,
short-circuit). If more current is required through the
EXTVCC switch than is specified, an external Schottky
diode can be interposed between the EXTVCC and INTVCC
pins. Ensure that EXTVCC ≤ VIN.
on, the switch node SW2 rises to V and the BOOST2
IN
pin rises to approximately V + INTV . When the bottom
IN
CC
MOSFET switch B turns on, the switch node SW2 drops
to low and the boost capacitor C is charged through D
B
B
from INTV . When the top MOSFET switch D turns on,
CC
the switch node SW1 rises to V
rises to approximately V
and the BOOST1 pin
CC
OUT
+ INTV . When the bottom
OUT
MOSFET switch C turns on, the switch node SW1 drops
to low and the boost capacitor C is charged through D
A
A
from INTV . The boost capacitors C and C need to
CC
A
B
store about 100 times the gate charge required by the top
MOSFET switch A and D. In most applications a 0.1μF to
0.47μF, X5R or X7R dielectric capacitor is adequate.
The following list summarizes the three possible connec-
Run Function
tions for EXTV :
CC
The RUN pin provides simple ON/OFF control for the
LTC3780. Driving the RUN pin above 1.5V permits the
controller to start operating. Pulling RUN below 1.5V puts
theLTC3780intolowcurrentshutdown.Donotapplymore
than 6V to the RUN pin.
1. EXTV left open (or grounded). This will cause INTV
CC
CC
to be powered from the internal 6V regulator at the cost
of a small efficiency penalty.
2. EXTV connected directly to V
(5.7V < V
< 7V).
CC
OUT
OUT
This is the normal connection for a 6V regulator and
provides the highest efficiency.
Soft-Start Function
Soft-start reduces the input power sources’ surge cur-
rents by gradually increasing the controller’s current
limit (proportional to an internally buffered and clamped
3. EXTV connected to an external supply. If an external
CC
supply is available in the 5.5V to 7V range, it may be
used to power EXTV provided it is compatible with
CC
equivalent of V ).
ITH
the MOSFET gate drive requirements.
An internal 1.2μA current source charges up the C
SS
Output Voltage
capacitor. As the voltage on SS increases from 0V to
2.4V, the internal current limit rises from 0V/R
to
SENSE
The LTC3780 output voltage is set by an external feedback
resistivedividercarefullyplacedacrosstheoutputcapacitor.
Theresultantfeedbacksignaliscomparedwiththeinternal
precision 0.800V voltage reference by the error amplifier.
The output voltage is given by the equation:
150mV/R
. The output current limit ramps up slowly,
SENSE
taking1.5s/μFtoreachfullcurrent.Theoutputcurrentthus
ramps up slowly, eliminating the starting surge current
required from the input power supply.
2.4V
1.2µA
T
=
•CSS = 1.5s/µF •C
SS
(
)
IRMP
R2
R1
⎛
⎝
⎞
VOUT = 0ꢀ8V s 1+
⎜
⎟
⎠
Do not apply more than 6V to the SS pin.
Topside MOSFET Driver Supply (C , D , C , D )
A
A
B
B
Current foldback is disabled during soft-start until the
voltage on C reaches 2V. Make sure C is large enough
SS
SS
Referring to Figure 11, the external bootstrap capacitors
when there is loading during start-up.
C and C connected to the BOOST1 and BOOST2 pins
A
B
3780fe
20
LTC3780
APPLICATIONS INFORMATION
The Standby Mode (STBYMD) Pin Function
Fault Conditions: Overvoltage Protection
Thestandbymode(STBYMD)pinprovidesseveralchoices
for start-up and standby operational modes. If the pin is
pulled to ground, the SS pin is internally pulled to ground,
preventing start-up and thereby providing a single control
pin for turning off the controller. If the pin is left open or
bypassedtogroundwithacapacitor,theSSpinisinternally
providedwithastartingcurrent,permittingexternalcontrol
for turning on the controller. If the pin is connected to a
A comparator monitors the output for overvoltage con-
ditions. The comparator (OV) detects overvoltage faults
greater than 7.5% above the nominal output voltage.
When the condition is sensed, switches A and C are
turned off, and switches B and D are turned on until the
overvoltage condition is cleared. During an overvoltage
condition, a negative current limit (V
= –60mV) is
SENSE
set to limit negative inductor current. When the sensed
currentinductorcurrentislowerthan–60mV,switchAand
C are turned on, and switch B and D are turned off until
the sensed current is higher than –20mV. If the output is
still in overvoltage condition, switch A and C are turned
off, and switch B and D are turned on again.
voltage greater than 1.25V, the internal regulator (INTV )
CC
will be on even when the controller is shut down (RUN
pin voltage < 1.5V). In this mode, the onboard 6V linear
regulator can provide power to keep-alive functions such
as a keyboard controller.
Fault Conditions: Current Limit and Current Foldback
Efficiency Considerations
The maximum inductor current is inherently limited in a
current mode controller by the maximum sense voltage.
In boost mode, maximum sense voltage and the sense
resistance determines the maximum allowed inductor
peak current, which is:
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in circuit produce losses, four main sources
account for most of the losses in LTC3780 circuits:
160mV
IL(MAX,BOOST)
=
RSENSE
2
1. DC I R losses. These arise from the resistances of the
MOSFETs, sensing resistor, inductor and PC board
traces and cause the efficiency to drop at high output
currents.
In buck mode, maximum sense voltage and the sense
resistance determines the maximum allowed inductor
valley current, which is:
130mV
RSENSE
2. Transition loss. This loss arises from the brief amount
of time switch A or switch C spends in the saturated
region during switch node transitions. It depends upon
the input voltage, load current, driver strength and
MOSFET capacitance, among other factors. The loss
is significant at input voltages above 20V and can be
estimated from:
IL(MAX,BUCK)
=
To further limit current in the event of a short circuit to
ground, the LTC3780 includes foldback current limiting.
If the output falls by more than 30%, then the maximum
sense voltage is progressively lowered to about one third
of its full value.
–1
Transition Loss ≈ 1.7A • V • I
• C
• f
IN2 OUT
RSS
where C
is the reverse transfer capacitance.
RSS
3780fe
21
LTC3780
APPLICATIONS INFORMATION
3. INTV current. This is the sum of the MOSFET driver
Thehighestvalueofripplecurrentoccursatthemaximum
input voltage. In boost mode, the ripple current is:
CC
and control currents. This loss can be reduced by sup-
plying INTV current through the EXTV pin from a
CC
CC
V
IN
⎛
V
IN
⎞
ΔIL,BOOST
=
s 1n
high efficiency source, such as an output derived boost
⎜
⎟
⎠
f sL ⎝ VOUT
network or alternate supply if available.
ΔIL,BOOST s100
4. C and C
loss. The input capacitor has the difficult
IN
OUT
IRIPPLE,BOOST
=
%
joboffilteringthelargeRMSinputcurrenttotheregula-
tor in buck mode. The output capacitor has the more
difficult job of filtering the large RMS output current
I
IN
The highest value of ripple current occurs at V = V /2.
IN
OUT
in boost mode. Both C and C
are required to have
IN
OUT
A 6.8μH inductor will produce 11% ripple in boost mode
(V = 6V) and 29% ripple in buck mode (V = 18V).
2
low ESR to minimize the AC I R loss and sufficient
capacitance to prevent the RMS current from causing
additional upstream losses in fuses or batteries.
IN
IN
The R
resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances.
SENSE
5. Other losses. Schottky diode D1 and D2 are respon-
sible for conduction losses during dead time and light
load conduction periods. Inductor core loss occurs
predominately at light loads. Switch C causes reverse
recovery current loss in boost mode.
2s160mV sV
IN(MIN)
RSENSE
=
2sIOUT(MAX,BOOST) sVOUT + ΔIL,BOOST sV
IN(MIN)
Select an R
of 10mΩ.
SENSE
Whenmakingadjustmentstoimproveefficiency, theinput
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased. If there is no change in input
current, then there is no change in efficiency.
Output voltage is 12V. Select R1 as 20k. R2 is:
VOUT •R1
R2 =
–R1
0.8
Select R2 as 280k. Both R1 and R2 should have a toler-
ance of no more than 1%.
Design Example
Asadesignexample,assumeV =5Vto18V(12Vnominal),
IN
Next, choose the MOSFET switches. A suitable choice is
V
OUT
= 12V (5%), I
= 5A and f = 400kHz.
OUT(MAX)
the Siliconix Si4840 (R
= 0.009Ω (at V = 6V),
DS(ON)
= 150pF, θ = 40°C/W).
GS
Set the PLLFLTR pin at 2.4V for 400kHz operation. The
inductance value is chosen first based on a 30% ripple
current assumption. In buck mode, the ripple current is:
C
RSS
JA
The maximum power dissipation of switch A occurs in
boost mode when switch A stays on all the time. Assum-
⎛
⎞
VOUT
f sL
VOUT
ing a junction temperature of T = 150°C with ρ
=
J
150°C
ΔIL,BUCK
=
s 1n
⎜
⎝
⎟
1.5, the power dissipation at V = 5V is:
V
IN
⎠
IN
2
12
5
⎛
⎞
⎠
ΔIL,BUCK s100
PA,BOOST
=
s5 s1ꢀ5s0.009=1.94W
⎜
⎝
⎟
IRIPPLE,BUCK
=
%
IOUT
3780fe
22
LTC3780
APPLICATIONS INFORMATION
Double-check the T in the MOSFET with 70°C ambient
C is chosen to filter the square current in buck mode. In
J
IN
temperature:
this mode, the maximum input current peak is:
29%
2
⎛
⎝
⎞
T = 70°C + 1.94W • 40°C/W = 147.6°C
J
IIN,PEAK(MAX,BUCK) = 5s 1+
= 5.7A
⎜
⎟
⎠
The maximum power dissipation of switch B occurs in
buckmode. AssumingajunctiontemperatureofT =80°C
J
A low ESR (10mΩ) capacitor is selected. Input voltage
ripple is 57mV (assuming ESR dominate ripple).
with ρ
= 1.2, the power dissipation at V = 18V is:
80°C
IN
18–12
•52 •1.2•0.009 = 90mW
C
is chosen to filter the square current in boost mode.
OUT
PB,BUCK
=
18
In this mode, the maximum output current peak is:
12
5
11%
2
⎛
⎝
⎞
Double-check the T in the MOSFET at 70°C ambient
J
IOUT,PEAK(MAX,BOOST)
=
s5s 1+
=10.6A
⎜
⎟
⎠
temperature:
T = 70°C + 0.09W • 40°C/W = 73.6°C
J
A low ESR (5mΩ) capacitor is suggested. This capacitor
will limit output voltage ripple to 53mV (assuming ESR
dominate ripple).
ThemaximumpowerdissipationofswitchCoccursinboost
mode.AssumingajunctiontemperatureofT =110°Cwith
J
ρ
= 1.4, the power dissipation at V = 5V is:
110°C
IN
PC Board Layout Checklist
12–5 •12
(
)
PC,BOOST
=
•52 •1.4•0.009
The basic PC board layout requires a dedicated ground
plane layer. Also, for high current, a multilayer board
provides heat sinking for power components.
52
5
+ 2•123 • •150p•400k =1.27W
5
• The ground plane layer should not have any traces and
it should be as close as possible to the layer with power
MOSFETs.
Double-check the T in the MOSFET at 70°C ambient
J
temperature:
T = 70°C + 1.08W • 40°C/W = 113°C
J
•
Place C , switch A, switch B and D1 in one com-
IN
pact area. Place C , switch C, switch D and D2 in
The maximum power dissipation of switch D occurs
in boost mode when its duty cycle is higher than 50%.
OUT
one compact area. One layout example is shown in
Figure 10.
Assuming a junction temperature of T = 100°C with
J
ρ
= 1.35, the power dissipation at V = 5V is:
100°C
IN
V
SW2
SW1
V
OUT
IN
2
5
12
12
5
⎛
⎞
⎠
L
D2
QD
PD,BOOST
=
s
s5 s1ꢀ35s0ꢀ009= 0ꢀ73W
⎜
⎝
⎟
QA
Double-check the T in the MOSFET at 70°C ambient
J
D1
temperature:
QB
QC
T = 70°C + 0.73W • 40°C/W = 99°C
J
C
C
OUT
IN
R
SENSE
LTC3780
CKT
GND
3780 F10
Figure 10. Switches Layout
3780fe
23
LTC3780
APPLICATIONS INFORMATION
• Use immediate vias to connect the components (in-
cluding the LTC3780’s SGND and PGND pins) to the
ground plane. Use several large vias for each power
component.
• Connect the top driver boost capacitor C closely to the
A
BOOST1 and SW1 pins. Connect the top driver boost
capacitor C closely to the BOOST2 and SW2 pins.
B
• Connect the input capacitors C and output capacitors
IN
• Use planes for V and V
to maintain good voltage
C
OUT
closely to the power MOSFETs. These capaci-
IN
OUT
filtering and to keep power losses low.
tors carry the MOSFET AC current in boost and buck
mode.
• Floodallunusedareasonalllayerswithcopper.Flooding
with copper will reduce the temperature rise of power
components. Connect the copper areas to any DC net
• Connect V
pin resistive dividers to the (+) termi-
OSENSE
nalsofC
andsignalground. AsmallV
bypass
OUT
OSENSE
(V or GND).
capacitormaybeconnectedcloselytotheLTC3780SGND
pin. The R2 connection should not be along the high
current or noise paths, such as the input capacitors.
IN
• Segregate the signal and power grounds. All small-
signal components should return to the SGND pin at
one point, which is then tied to the PGND pin close to
the sources of switch B and switch C.
–
+
• RouteSENSE andSENSE leadstogetherwithminimum
PC trace spacing. Avoid sense lines pass through noisy
area,suchasswitchnodes.Thefiltercapacitorbetween
• Place switch B and switch C as close to the controller
as possible, keeping the PGND, BG and SW traces
short.
+
–
SENSE and SENSE should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the SENSE resistor. One layout example
is shown in Figure 12.
• Keep the high dV/dT SW1, SW2, BOOST1, BOOST2,
TG1 and TG2 nodes away from sensitive small-signal
nodes.
• Connect the I pin compensation network close to the
TH
IC, between I and the signal ground pins. The capaci-
TH
• The path formed by switch A, switch B, D1 and the C
tor helps to filter the effects of PCB noise and output
IN
capacitor should have shortleads andPCtracelengths.
voltage ripple voltage from the compensation loop.
The path formed by switch C, switch D, D2 and the
• ConnecttheINTV bypasscapacitor, C , closetothe
CC
VCC
C
capacitor also should have short leads and PC
OUT
trace lengths.
IC,betweentheINTV andthepowergroundpins.This
CC
capacitor carries the MOSFET drivers’ current peaks.
• Theoutputcapacitor(–)terminalsshouldbeconnected
as close as possible the (–) terminals of the input
capacitor.
Anadditional1μFceramiccapacitorplacedimmediately
next to the INTV and PGND pins can help improve
CC
noise performance substantially.
3780fe
24
LTC3780
APPLICATIONS INFORMATION
V
OUT
R
PU
V
C
PULLUP
OUT
C
A
1
2
24
23
C
SS
PGOOD BOOST1
SS
TG1
D
C
D2
LTC3780
+
D
A
3
4
22
SENSE
SENSE
SW1
C
C2
C
–
C
C
21
20
19
18
17
16
15
F
C
V
IN
C1
R
R
R
C
5
6
I
EXTV
TH
CC
CC
R2
VCC
L
R1
V
INTV
OSENSE
7
SGND
RUN
BG1
PGND
BG2
R
SENSE
8
D1
9
FCB
B
A
10
PLLFLTR
SW2
D
B
11
12
14
13
f
PLLIN
TG2
IN
C
B
C
IN
STBYMD BOOST2
R
IN
V
IN
3780 F11
Figure 11. LTC3780 Layout Diagram
PGND
C
R
R
SGND
3780 F12
Figure 12. Sense Lines Layout
3780fe
25
LTC3780
PACKAGE DESCRIPTION
G Package
24-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
7.90 – 8.50*
(.311 – .335)
1.25 p0.12
24 23 22 21 20 19 18 17 16 15 14
13
7.8 – 8.2
5.3 – 5.7
7.40 – 8.20
(.291 – .323)
0.42 p0.03
0.65 BSC
RECOMMENDED SOLDER PAD LAYOUT
5
7
8
1
2
3
4
6
9 10 11 12
2.0
5.00 – 5.60**
(.197 – .221)
(.079)
MAX
0o – 8o
0.65
(.0256)
BSC
0.09 – 0.25
0.55 – 0.95
(.0035 – .010)
(.022 – .037)
0.05
0.22 – 0.38
(.009 – .015)
TYP
(.002)
NOTE:
MIN
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
G24 SSOP 0204
3. DRAWING NOT TO SCALE
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
3780fe
26
LTC3780
PACKAGE DESCRIPTION
UH Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693 Rev D)
0.70 p0.05
5.50 p0.05
4.10 p0.05
3.45 p 0.05
3.50 REF
(4 SIDES)
3.45 p 0.05
PACKAGE OUTLINE
0.25 p 0.05
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
BOTTOM VIEW—EXPOSED PAD
PIN 1 NOTCH R = 0.30 TYP
OR 0.35 s 45o CHAMFER
R = 0.05
TYP
0.00 – 0.05
R = 0.115
TYP
0.75 p 0.05
5.00 p 0.10
(4 SIDES)
31 32
0.40 p 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
3.45 p 0.10
3.50 REF
(4-SIDES)
3.45 p 0.10
(UH32) QFN 0406 REV D
0.200 REF
0.25 p 0.05
0.50 BSC
NOTE:
1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3780fe
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC3780
TYPICAL APPLICATION
V
12V
5A
OUT
R
PU
22μF
16V, X7R
s 3
C
A
V
PULLUP
0.22μF
+
C
C
OUT
SS
1
24
23
330μF
16V
0.022μF
PGOOD BOOST1
2
D
SS
TG1
D2
Si7884DP
C
C2
LTC3780
+
B320A
D
A
47pF
BO540W
68pF
3
4
5
6
22
21
20
19
SENSE
SENSE
SW1
C
C1
C 0.1μF
F
R
–
C
C
V
0.01μF
IN
100k
Si7884DP
L
I
EXTV
4.7μH
TH
CC
CC
C
4.7μF
9mΩ
VCC
R1
8.06k, 1%
V
INTV
OSENSE
R2 113k, 1%
ON/OFF
7
8
18
17
16
15
SGND
RUN
BG1
PGND
BG2
9
B
D1
B340A
FCB
Si7884DP
10
PLLFLTR
SW2
D
B
10k
BO540W
C
22μF
35V
IN
A
11
12
14
13
+
PLLIN
TG2
Si7884DP
STBYMD BOOST2
2V
3.3μF
C
STBYMD
C 0.22μF
B
50V, X5R
10Ω
0.01μF
s 3
V
IN
100Ω
5V TO 32V
3780 TA02
100Ω
Figure 13. LTC3780 12V/5A, Buck-Boost Regulator
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
No R ™, 2.5V ≤ V ≤ 36V Burst Mode Operation, MSOP-10
LTC1871/LTC1871-1 SEPIC, Boost, Flyback Controller
LTC1871-7
SENSE
IN
Package
LTC3443
1.2A I , 600kHz, Synchronous Buck-Boost DC/DC
V : 2.4V to 5.5V, V : 2.4V to 5.25V, I = 28μA, I < 1μA,
OUT
IN
OUT
Q
SD
Converter
MS Package
LTC3444
500mA I , 1.5MHz Synchronous Buck-Boost DC/DC
V : 2.7V to 5.5V, V : 0.5V to 5.25V, Optimized for WCDMA RF
OUT
IN
OUT
Converter
Amplifier Bias
LTC3531/LTC3531-3 200mA I , Synchronous Buck-Boost DC/DC Converter
V : 1.8V to 5.5V, V : 2V to 5V, I = 35μA, I < 1μA,
OUT
IN
OUT
Q
SD
LTC3531-3.3
MS, DFN Packages
LTC3532
500mA I , 2MHz, Synchronous Buck-Boost DC/DC
V : 2.4V to 5.5V, V : 2.4V to 5.25V, I = 35μA, I < 1μA,
OUT
IN
OUT
Q
SD
Converter
MS, DFN Packages
LTC3533
2A Wide Input Voltage Synchronous Buck-Boost DC/DC
Converter
V : 1.8V to 5.5V, V : 1.8V to 5.25V, I = 40μA, I < 1μA,
IN
OUT
Q
SD
DFN Package
LTC3785/LTC3785-1 10V, High Efficiency, Synchronous, No R , Buck-Boost
SENSE
Controller
V : 2.7V to 10V, V : 2.7V to 10V, I = 86mA, I < 15μA,
IN
OUT
Q
SD
QFN-24 Package
LTC4444/LTC4444-5 High Voltage Synchronous N-Channel MOSFET Driver
V
IN
up to 100V, Used with the LTC3780 for Higher V Applications
IN
LTM4605
5A to 12A Buck-Boost μModule™
4.5V ≤ V ≤ 20V, 0.8V ≤ V
≤ 16V, 15mm × 15mm × 2.8mm
IN
OUT
LGA Package
LTM4607
5A to 12A Buck-Boost μModule
4.5V ≤ V ≤ 36V, 0.8V ≤ V
≤ 24V, 15mm × 15mm × 2.8mm
IN
OUT
LGA Package
No R
and μModule are trademarks of Linear Technology Corporation
SENSE
3780fe
LT 0309 REV E • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
28
●
●
© LINEAR TECHNOLOGY CORPORATION 2005
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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